Method and system for a polarization mode dispersion tolerant optical homodyne detection system with optimized transmission modulation

ABSTRACT

An optical homodyne communication system and method in which a side carrier is transmitted along with data bands in an optical data signal, and upon reception, the side carrier is boosted, shifted to the center of the data bands, and its polarization state is matched to the polarization state of the respective data bands to compensate for polarization mode dispersion during transmission. By shifting a boosted side carrier to the center of the data bands, and by simultaneously compensating for the effects of polarization mode dispersion, the provided system and method simulate the advantages of homodyne reception using a local oscillator. The deleterious effects of chromatic dispersion on the data signals within the data bands are also compensated for by applying a corrective function to the data signals which precisely counteracts the effects of chromatic dispersion.

CROSS-REFERENCES TO RELATED APPLICATIONS

This application is a divisional of U.S. patent application Ser. No.12/554,241 filed Sep. 4, 2009, which is divisional of U.S. patentapplication Ser. No. 09/871,216 filed on May 31, 2001, which is relatedto copending and commonly assigned U.S. patent application Ser. No.09/748,750 , filed in the United States Patent and Trademark office onDec. 26, 2000, entitled “Method, System and Apparatus for OpticallyTransferring Information”, which is expressly incorporated herein in itsentirety by reference thereto.

FIELD OF THE INVENTION

The present invention relates to optical data communication, and inparticular, relates to a method and optical data communication systemthat improves signal-to-noise ratio of optical data signals, counteractspolarization mode dispersion and improves robustness to fibernonlinearities.

BACKGROUND INFORMATION

Currently, optical data communication systems are being upgraded from a10 Gb/s data transmission rate up to a 40 Gb/s transmission rate.However, data transmission at 40 Gb/s (or higher) presents extensivedesign challenges because the effects of polarization mode dispersion(PMD), chromatic dispersion and fiber non-linear effects such ascross-phase modulation become more dominant at the higher transmissionrates. In particular, the limit of tolerable polarization modedispersion, usually defined as 14% of the data bit duration, is only 3.5ps at a 40 Gb/s transmission rate. A 3.5 ps polarization mode dispersiontranslates to an attainable reach of several hundred kilometers oversingle mode fiber which has a typical fiber PMD of 0.1 ps/km^(1/2).

Current optical communications systems, such as the PMD compensationarrangement described in U.S. Pat. No. 6,130,766 to Cao, generallyattempt to compensate for PMD by splitting received optical signals intox and y mode components having orthogonal polarization, and thenadjusting the delay on one of the orthogonal components to align themodes. This arrangement requires significant signal processing anddifferential delays to cover the range of frequencies carrying data.

Nonlinearities induced during optical transmission are also amplified athigher data rates. While it is necessary for accurate detection thatoptical data signals be at least 20dB above background noise, if thedata signals are transmitted with too much power, nonlinearities canplay a greater role in distorting the signal. In addition, in coherentsystems typical heterodyne optical reception systems suffer an inherent3dB penalty with respect to homodyne systems and introduce phase noisethrough use of a local oscillator, and thereby add a further level ofcomplexity and constraints to optical system design.

What is therefore needed is a cost-effective method and system thatcompensates for PMD, optimizes SNR performance and minimizes phase noiseand nonlinearities associated with transmission over fiber at high datatransmission rates.

SUMMARY OF THE INVENTION

The present invention meets the above objectives by providing an opticalhomodyne communication system and method in which a reduced amplitudeside carrier is transmitted along with data bands in an optical datasignal, and upon reception, the side carrier is boosted, shifted to thecenter of the data bands, and its polarization state is matched to thepolarization state of the respective data bands to compensate forpolarization mode dispersion during transmission. This scheme achievesthe signal-to-noise benefits of homodyne reception without incurring theconventional restrictions and complications of homodyne reception suchas requiring the phase of a signal from a local oscillator to be lockedto the phase of the optical signal.

According to one embodiment, the present invention provides a method ofoptical communication that begins with providing a quadrature modulatedoptical data including two data bands separated in frequency, each databand having in-phase and quadrature components. The power of thequadrature modulated optical data signal is limited in order to limitnon-linear effects by reducing the power of the optical data signalduring transitional states in which data symbols transmitted in theoptical data signal change in value, and in particular by reducing thepower to zero such that transmitted power decreases to zero atapproximately the mid point of the transitional states. The optical datasignal is combined with a side carrier at a single frequency between thetwo data bands of the optical data signal and then transmitted acrossoptical fiber to a receiver.

At the receiver, the side carrier is separated from the two data bandsof the combined optical data signal and increased in amplitude relativeto the data. The side carriers are then shifted to the middle of each ofthe respective two data bands. Since the relationship between thepolarization state of the side carriers and the polarization state ofthe data bands does not stay constant during transmission over opticalfiber, the polarization state of the shifted side carriers is adjustedto match the polarization state of the data bands at which they arecentered.

The present invention further provides a method of compensating for theeffects of chromatic dispersion during transmission over optical fiberby separating the in-phase and quadrature components of the two databands prior to optoelectric conversion, and, after optoelectricconversion, compensating for chromatic dispersion by applying acorrective function to each of the in-phase and quadrature components ofthe data bands, the corrective function precisely counteracting theeffects of chromatic dispersion on the in-phase and quadraturecomponents.

The present invention also provides a method of providing informationconcerning a transmission device by providing an optical data signalhaving data bands and a side carrier with the side carrier modulated tocarry an identification code, the identification code includinginformation concerning the transmitter. According to an embodiment ofthe present invention, the information concerning a transmitter embeddedin the side carrier includes parameters used in the corrective functionto precisely counteract the effects of chromatic dispersion.

An optical data signal transmitter is provided for generating thequadrature modulated optical data signal including at least one sidecarrier. The transmitter includes a Mach-Zender modulator whichgenerates an optical carrier signal by modulating a pair of sidecarriers onto an input optical signal. The optical carrier signal ismodulated by at least two phase modulators which modulate a pair of datasignals, in quadrature, onto the optical carrier signal, outputting anoptical data signal including at least two data bands. By spreading thedata bands onto the pair of side carriers, the amplitude of the opticaldata signal is reduced to zero during transitions between data symbols.The transmitter also includes a second Mach-Zender modulator whichimprints a low-frequency TX ID (transmitter identification code) sidecarrier onto the input optical signal. The TX ID signal side carrierthen combined with the optical data signal for transmission. Thetransmitted identification code includes information concerning thetransmitter, such as its location, from which the distance between thetransmitter and a receiver may be deduced.

The present invention further provides a receiver for implementinghomodyne reception. The receiver includes a side carrier boosting modulefor increasing the amplitude of the side carrier relative to the databands in the optical data signal. The receiver further includes a sidecarrier shifting module coupled to the side carrier boosting modulewhich shifts the side carrier into two shifted carriers. Each of theshifted carriers is shifted to the center of one of the data bands. Inaddition, means for compensating polarization mode dispersion that arecoupled to the side carrier shifting module match the polarizationstates of the shifted carriers to the data bands by adjusting either thepolarization state of the shifted carriers or the polarization state ofthe data bands. After optoelectric conversion of the optical datasignal, the receiver employs a chromatic dispersion correction stagethat includes circuits that apply transfer functions to the in-phase andquadrature detected data channels

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of a transmitter according to an embodiment ofthe present invention.

FIG. 2 a shows the spectrum of an optical carrier signal at the outputof MZ1 of FIG. 1 according to an embodiment of the present invention.

FIG. 2 b shows the spectrum of an optical data signal at the output ofQMZ3 of FIG. 1 after data modulation in quadrature according to anembodiment of the present invention.

FIG. 2 c shows the spectrum of an optical data signal at the output ofQMZ2 of FIG. 1 according to an embodiment of the present invention.

FIG. 3 shows a 10G symbol per second Quadrature Return to Zero (QRZ)constellation diagram of the output from QMZ2 and QMZ3.

FIG. 4 a shows the spectrum of an optical data signal at the output ofC1 of FIG. 1 according to an embodiment of the present invention.

FIG. 4 b shows the spectrum of an optical data signal at the output ofC2 of FIG. 1 according to an embodiment of the present invention.

FIG. 4 c shows the spectrum of an optical data signal at the output ofthe DWDM of FIG. 1 according to an embodiment of the present invention.

FIG. 5 is a block diagram of a receiver according to an embodiment ofthe present invention.

FIG. 6 is a block diagram of a first embodiment of the side carrierboosting module according to the present invention.

FIG. 7 is a block diagram of a second embodiment of the side carrierboosting module according to the present invention which employs theStimulated Brillouin Scattering (SBS) effect.

FIG. 8 a shows the spectrum of an optical carrier signal at the outputof the FP4 of FIG. 5 according to an embodiment of the presentinvention.

FIG. 8 b shows the spectrum of an optical carrier signal at the outputof the S7 of FIG. 5 according to an embodiment of the present invention.

FIG. 8 c shows the spectrum of an optical carrier signal at the outputof the PBS3 of FIG. 5 according to an embodiment of the presentinvention.

FIG. 8 d shows the spectrum of an optical carrier signal at the outputof the SS1 of FIG. 5 according to an embodiment of the presentinvention.

FIG. 8 e shows the spectrum of an optical carrier signal at the outputof the C5 of FIG. 5 according to an embodiment of the present invention.

FIG. 8 f shows the spectrum of an optical carrier signal at the outputof the C6 of FIG. 5 according to an embodiment of the present invention.

FIG. 9 is a block diagram of a chromatic dispersion compensation circuitaccording to an embodiment of the present invention.

FIG. 10 a is a block diagram of a microstrip implementation of a circuitthat applies a COS transfer function to an input signal according to anembodiment of the present invention.

FIG. 10 b is a block diagram of a microstrip implementation of a circuitthat applies a SIN transfer function to an input signal according to anembodiment of the present invention.

DETAILED DESCRIPTION

I. Transmission

In accordance with the present invention, at a transmitter, a pair ofside carriers is modulated onto each side of a monochromatic opticalcarrier signal, which is then split into two channels. Each opticalcarrier signal channel is modulated with two 10 Gb/s data signals in anorthogonal phase relationship to one another. The data signals arespread onto the two side carriers in each channel, and in effect, arespread out by fifty percent in the frequency domain. This spreading isequivalent to multiplication by a sine wave at half the data rate, andresults in each data symbol returning to zero between transitions,referred to as quadrature-return-to-zero (QRZ). Using QRZ, the power ofthe optical data signal is made independent of the data pattern. Thepolarization of one of the optical data signal channels is then shifted,and one of the channels is combined with a channel of the originalmonochromatic carrier that has been modulated with a transmissionidenfitication carrier of less than 100 kHz.

The two optical data signal bands, which each carry a 20 Gb/s datastream, are combined and either multiplexed with adjacent channels atsimilar frequency and orthogonal polarization or one of the two channelsis shifted in polarization to match the other channel. In either case,the optical data signals are multiplexed according to a Dense WaveDivision Mulitplexing (DWDM) scheme and transmitted along long haulfiber to a destination receiver.

FIG. 1 illustrates an embodiment of a transmitter according to thepresent invention, which may be implemented on a Lithium-Niobate chip,for example. An optical signal generator SG1, which may be a laser,generates a monochromatic, polarized optical carrier at a referencefrequency which for purposes of the following discussion is designatedas the origin (0 GHz) in terms of relative frequency. The optical signalis thereafter split into two channels, an upper channel going toMach-Zender modulator MZ1 and a lower channel being transmitted toMach-Zender modulator MZ4. The division of light intensity between thetwo channels can be uneven with the lower channel receiving, forexample, just 10 percent of the light intensity generated by SG1. Atnarrow-band modulator MZ4, the lower channel of the optical signal ismodulated with a “Transmitter Identification” (TX ID) tone in thefrequency range of 10 KHz to 100 KHz above the reference frequency. Atmodulator MZ1, two sets of side carriers at +/−20 GHz and +/−30 GHz aremodulated onto the optical signal. The spectrum for the modulatedsignal, denoted an optical carrier signal, is shown in FIG. 2 a. Asshown in the figure, the resulting spectrum output from MZ1 shows fourpeaks, two below the reference frequency at −30 GHz and −20 GHz relativeto the reference frequency, and two above the reference frequency at +20GHz and +30 GHz.

The output of modulator MZ1 is further split into an upper channel whichis transmitted to quadrature data modulator QMZ3 and a lower channelwhich is transmitted to quadrature data modulator QMZ2. Data modulatorQMZ2 imprints two individual 10 Gb/s data streams in quadrature (inorthogonal phase relationship) CH.1 and CH.2 onto each of the pairs ofside carriers above and below the reference frequency. Similarly, datamodulator QMZ3 imprints individual 10 Gb/s data streams CH.3 and CH.4onto each of the pairs of side carriers in the optical carrier signal.Respective bias control electrodes VB2 and VB3 assist in keeping thedata streams in quadrature. Spectra of the outputs from QMZ3 and QMZ2are shown in FIG. 2 b and FIG. 2 c respectively. As can be discerned inFIG. 2 b and FIG. 2 c, the output spectra from QMZ3 and QMZ2 show twodata bands, one extending from −40 GHz to −10 GHz and another extendingfrom +10 GHz to +40 GHz relative to the reference frequency.

By imprinting two 10 Gb/s data streams in quadrature, in effect, 20 Gb/sof data are modulated onto each pair of side carriers (−30, −20 GHz and+20, +30 GHz, respectively) and each 20 Gb/s data band covers 30 GHz inthe frequency domain. By providing two side carriers, with one sidecarrier in the pair a clock rate away from the other (i.e., 30 GHz beinga clock away from 20 GHz), the data bits in both I and Q format aremultiplied in the time domain by a 5 GHz sinusoid which crosses zeroevery 100 ps. Thus, the total data signal always crosses through zero inbetween any pair of symbols (any pair of I,Q data), referred to asquadrature-return-to-zero (QRZ) modulation.

FIG. 3 illustrates the key property of the QRZ format, showing that thetrajectory between two successive symbols always leads through the I-Qorigin. Each corner of the figure represents a pair of I, Q data symbols(e.g., I=1,Q=−1 or I=−1, Q=1). As shown, to get from adjacent cornerpoints I=1, Q=1 (upper right corner) to I=1, Q=−1 (lower right corner)the optical data signal must travel through the origin (0,0). Duringeach trajectory through the origin, the power of the signal, which isproportion to the square of its amplitude, goes to zero.

Returning to FIG. 1, the output from modulator QMZ3 is input to apolarization transformer PT1, which shifts the polarization of theoptical data signal output from QMZ3 90 degrees. The polarization of thesignal output from PT1 is arbitrarily illustrated by parallel lines asparallel polarization as opposed to a perpendicular polarization of theoriginal optical signal. Furthermore, the output optical data signalfrom modulator QMZ2 is combined at combiner C1 with the TX ID pilotsignal from MZ4. The output from C1 is shown in FIG. 4 a. As notedabove, the intensity of the TX ID signal is reduced in comparison withthe optical data signal from QMZ2. It is also noted that thepolarization of the output signal from C1 is shown as perpendicular,since the polarization of the output from C1 remains unchanged from theoriginal polarization. Thereafter, the output signal from PT1 iscombined with the output signal from combiner C1 at C2. The spectrum ofthe output signal out of C2 is shown in FIG. 4 b. As can be discerned,the spectrum includes data channels 1, 2, 3 and 4 in both lower andupper data bands. Channels 1 and 2 are in perpendicular polarization andchannels 3 and 4 are in parallel polarization. The reference carrier atapproximately 0 GHz from MZ4 is in perpendicular polarization.

According to the illustrated embodiment, the output signal from C2 isinput to a polarization beam splitter PBS1 which splits the signal intoperpendicular and parallel polarized components, thereby separating thedata channels 1 and 2 from channels 3 and 4. The perpendicular component(containing data channels 1 and 2 as well as the central referencefrequency) is transmitted along lower path 102 to a first channel of adense wave division multiplexer DWDM, the parallel component (containingdata channels 3 and 4) is input to a polarization transformer PT2, whichrotates the polarization of the parallel component back into aperpendicular state. The output from PT2 is then input to a second DWDMchannel. Each DWDM channel acts as a band pass filter and passes onlyfrequencies that fall within a 50 GHz band. Assuming for illustrativepurposes that DWDM channel 1 passes frequencies from −50 GHz to 0 GHzrelative to the reference frequency, and DWDM channel 2 passesfrequencies from 0 to +50 GHz, data channels 1 and 2 are passed only inthe data band from −40 GHz to −10 GHz and while data channels 3 and 4are passed only in the data band from +10 GHz to +40 GHz. The DWDMmultiplexes each of the passed bands onto a long haul fiber (not shown).The output spectrum from −50 GHz to +50 GHz output from the DWDM isshown in FIG. 4 c. The adjacent DWDM channels each pass 20 Gb/s of data,combining for a total of 40 Gb/s.

In an alternative embodiment, a polarization multiplexing scheme may beused, making it unnecessary to separate data channels 1 and 2 from datachannels 3 and 4. As described in related and commonly owned application[Ser. No. 09/782,354] hereby incorporated for reference, the pairs ofdata channels can occupy the same data band if their polarization statesremain orthogonal and thus do not interfere with each other. In thisimplementation, the polarization beam splitter PBS1 is not needed andthe output from C2 can be sent directly to one of the DWDM inputchannels.

II. Reception

In accordance with the present invention, a homodyne reception system isemployed to receive the optical data signal generated as describedabove. Upon reception, the transmitted side carrier at the referencefrequency is boosted to increase the signal-to-noise ratio (SNR) of theoptical data signal and to compensate for the attenuation of the sidecarrier in the transmitter. The boosting of the side carrier increasesthe SNR because of the implementation of homodyne reception in whichoverall detected signal power is increased in proportion to the power ofthe local oscillator, or in the present case (as will be discussedbelow), the transmitted side carrier.

Once the amplitude of the side carrier power is boosted relative to thetransmitted data bands, the side carrier is shifted by +/−25 GHz intotwo side carriers that are each shifted to the center of one of the twodata bands to further implement homodyne reception.

After the shifting of the side carriers, the two side carriers areseparated and then modified by polarization controllers which match thetime-varying polarization state of each the side carriers to thedifferent time-varying polarization state of the respective data bands,thus overcoming the effects of polarization mode dispersion bycontrolling the polarization at only a single frequency.

According to an embodiment of the present invention, a chromaticdispersion compensation stage is used to counter the effects ofdispersion during transmission over long haul fiber. Since the effectsof dispersion can be modeled as a transfer function that is applied tothe I and Q data signals, the chromatic dispersion compensation stageapplies a compensating correction function that effectively counteractsthe transfer function, rendering the I and Q data signals into theiroriginal non-dispersed state.

FIG. 5 illustrates an embodiment of a homodyne receiver according to thepresent invention. An optical data signal is received first by a sidecarrier boosting module 200 for which the present invention provides twoexemplary embodiments. In a first embodiment of the side carrierboosting module, shown in FIG. 6, the optical data signal is first inputto an optical amplifier EDFA1, which may be, for example, anerbium-doped fiber amplifier (EDFA). It is noted that all furtheroptical amplifiers used in the implementations described below may beimplemented as erbium-doped fiber amplifiers. The optical amplifierEDFA1 amplifies the entire spectrum of the received signal by, forexample, approximately 15-18 dB. The amplified signal output from EDFA1is split at S3 between an upper branch that is coupled to a Fabry-Perotresonator FP1 and a lower branch that is coupled to an attenuator ATT1.

The Fabry Perot resonator FP1 functions as a high-Q filter that nearlycompletely filters out all frequencies excepts for a series offrequencies that are separated by, for example, 100 Ghz which, accordingto the International Telecommunication Union (ITU) grid, is the amountof bandwidth allocated for each channel. The resonator FP1 is adjustedto pass the side carrier at the reference frequency and filter out thedata bands of the optical data signal. It is noted in this regard thatit is contemplated that the embodiments of the present invention be usedin the context of the ITU grid, and that the reception approachdescribed allows for simultaneous processing of side carriers for aplurality of ITU grid-spaced channels. The lower branch passed to ATT1,which contains both the data bands and the side carrier is attenuated.The signals output from FP1 and ATT1 are combined in combiner C4 andthen passed to a further optical amplifier EDFA2 where the combinedsignal is again amplified by, for example, approximately 15-18 dB.Because the side carrier was isolated and boosted in FP1 and the databands were attenuated in ATT1, the combined signal contains a sidecarrier boosted at least 10 dB in amplitude relative to the data bands.

A second embodiment of the side carrier boosting module, whichadvantageously makes use of the amplitude-enhancing effect of StimulatedBrillouin Scattering (SBS), is shown in FIG. 7. The SBS effect causes afirst optical signal having narrow frequency band around frequency X tobe amplified when collides with a signal of frequency X+≈11 GHztraveling in the opposite direction. Referring to FIG. 7, the receivedsignal is input to optical amplifier EDFA3 which amplifies the entirespectrum of the input signal. The signal output from amplifier EDFA3 istransmitted to optical isolator ISL1, which permits optical signal totravel only in one direction (the direction indicated by the arrow inthe figure) and prevents optical signals being reflected or transmittedback toward the amplifier EDFA3. From the optical isolator ISL1, theoptical data signal is split into two branches at splitter S4.

A first upper branch from splitter S4 leads to Fabry Perot resonatorFP2, which passes the side carrier (and other modes in the series offrequencies) in between the data bands. FP controller 1 automaticallyadjusts the resonator FP2 so that it correctly passes the side carrierusing input from splitter S5 and filters out the data bands. The outputfrom FP2 is delivered to external modulator XMOD 1, which also receivesan 11 GHz signal from a 11 GHz oscillator through an 11 GHz amplifier.The external modulator XMOD 1 modulates the 11 GHz signal onto the sidecarrier. The spectrum of the output from the modulator XMOD 1 thereaftercontains the reference frequency and two side frequencies located 11 GHzboth above and below the reference frequency. This output signal is thentransmitted to another resonator FP3, which is adjusted by FP controller2 to center on (and pass) only the side frequency 11 GHz above thereference side carrier frequency. The resulting signal, carryingsubstantially a single frequency at the reference frequency +11 GHz, isamplified in optical amplifier EDFA4 and then input to circulator CIRC1. The circulator passes signals in a counter-clockwise direction. Morespecifically, CIRC 1 passes the output from EDFA4 leftwards in acounter-clockwise rotation towards the output of optical isolator ISL 2.It is noted that the side carrier boosting scheme is also intended beused in conjunction with a dense wave division multiplexing scheme.Thus, the side carrier boosting module can simultaneously process andboost a plurality of side carriers spaced in frequency according to ITUchannel spacing.

Simultaneously, the optical signal in the lower branch from splitter S4is transmitted through isolator ISL 2 and then meets with the opticalsignal from the upper branch output from the circulator CIRC 1. Thiscollision of the two optical signals traveling in opposite directionsgenerates the SBS non-linear effect. According to one implementation,the fiber connecting isolator ISL 2 and circulator CIRC 1 can bedispersion compensating fiber which, due to its relatively smallercross-section, promotes higher intensity and more pronounced non-lineareffects such as SBS. When the optical data signal containing thereference side carrier collides with the 11 GHz side frequency signalfrom CIRC 1, a narrow band including the side carrier in the opticaldata signal is amplified relative to the data bands due to the SBSeffect as explained above. This modified optical data signal thenreaches the circulator CIRC 1 from which it passes in thecounter-clockwise direction to optical amplifier EDFA5, which amplifiesthe entire spectrum of the modified optical data signal by 15-18 dB. Theoutput from EDFA5 is the final output of the second embodiment of theside carrier boosting module 200.

Returning to FIG. 5, the optical data signal output from the sidecarrier boosting module 200 is input to circulator CIRC 2, which in turntransmits the signal in a counter-clockwise direction to Fabry-Perotresonator FP4, having a free spectral range (FSR) of 100 GHz and finesseon the order of 1000. The resonator FP4 is also tuned to select the sidecarrier at (approximately) the reference frequency (0 GHz). FIG. 8 ashows a spectrum of the signal output from FP4, indicating that the databands have again been filtered out. The data bands that are filtered outat FP4 are resent back toward circulator CIRC 2, where they areredirected in a counter-clockwise direction towards splitter S7. Thespectrum of the output from splitter S7, which includes the two filtereddata bands at −40 GHz to −10 GHz and +10 GHz to +40 GHz, is shown inFIG. 8 b.

It is noted that when the optical data signal is transmitted over longhaul fiber between the transmitter and the receiver, the polarizationstate of the transmitted signal is scrambled, with the result that thereceived signal has an unknown time-varying polarization state. Sincethe time-varying polarization state varies with frequency, the sidecarrier is expected to have a different time-varying polarization statethan either of the data bands because it is separated from the centersof data bands by 25 GHz. When the output from resonator FP4 is fed tothe side carrier shifting module 210, the side carrier's orthogonalpolarization states are split in polarization beam splitter PBS2, andthen each of the orthogonal signals are separately modulated by 25 GHzin XMOD 2 and XMOD 3, respectively, and then joined back in PBS3. Theoutput from PBS3 is illustrated in FIG. 8 c, which shows two sidecarriers at −25 GHz and +25 GHz from the reference frequency,respectively. The output from PBS2 is passed on to Fabry-Perot filterFP5 (FSR=50 GHz, finesse>500) which passes both the 25 GHz left andright shifted side carriers, and transmits them to circulator CIRC 3.Circulator CIRC 3 delivers shifted SC's to reflective polarizationcontrollers PC 1, PC 2, through respective adjacent 50 GHz-spacedchannels of WDM demultiplexer DWDM 2. The polarization controllers PC 1,PC 2 are constructed to provide control of the phase of the signalsreflected from the polarization controllers back to the demultiplexerDWDM 2. Such control may be used, for instance, in order to compensatefor the effective fiber length between the polarization controllers PC1, PC 2 and the demultiplexer DWDM 2. In one implementation, thepolarization controllers PC 1, PC 2 include mirrors and piezoelectricactuators to adjust the distance the reflected signal travels, which inturn controls the phase of the reflected optical signal.

Each polarization controller PC 1, PC 2 is used to transform thetime-varying polarization state of one of the two side carriers so thatthe polarization states of each side carrier matches the time varyingpolarization state of the respective data bands which are centered atthe side carrier (−25 GHz and +25 GHz). To accomplish this, eachpolarization controller PC 1, PC2 obtains feedback from the photodiodesthat receive the data bands. PC 1 receives the feeback via bias-Tcouplers BT 1 and BT 3, while PC 2 receives feedback via bias-T couplersBT 2 and BT 4. As will be described below, the demultiplexers at the topof FIG. 5, DWDM 3, DWDM 4, receive both the data bands and the sidecarriers, filter them into separate, adjacent frequency channels andthen effectively multiply the side carrier and data bands together atphotodiodes PD1, PD2, PD3 and PD4 (and other photodiodes of adjacentchannels that are not shown) which respond to the intensity of thesignal (i.e., the square of the amplitude). The product signal outputfrom the photodiodes is delivered to the respective polarizationcontrollers PC 1, PC 2 via bias-T couplers BT 1, BT 2, BT 3 and BT 4.The outputs from BT 1 and BT 3, which contain converted data signals 1and 2, corresponding to data channels 1 and 2, are combined to providefeedback to polarization controller PC 1, and the outputs from BT 2 andBT 4, which contain data signals 3 and 4, corresponding to data channels3 and 4, are combined to provide feedback polarization controller PC 2.It is noted that the data signals 1 and 2 are expected to have a similarpolarization state since, during transmission, they occupy the samefrequency range. Equally, data signals 3 and 4, corresponding to datachannels 3 and 4, are expected to have a similar polarization state. Atthe polarization controllers PC 1, PC 2, the time-varying polarizationof the combined product signals are compared to the polarization stateof the individual side carrier signals.

By continually adjusting the polarization of the side carrier signal andthen comparing the modified polarization state to the combined productsignals, the polarization controllers PC 1, PC 2 can accurately matchthe time-varying polarization state of each of the side carriers withthe time-varying polarization state of the corresponding data bands.This technique takes advantage of fact that it is easier to adjust thesingle polarization state of a single side carrier frequency than toadjust the multitude of polarization states of a band of frequencies,for example, a 20 GHz data band, via wide-band polarizationcompensation. However, polarization mode dispersion compensation canalso be performed here by adjusting the average polarization of the databand, which is treated as having a single polarization, and thenmatching to the polarization of the side carrier.

Returning once again to FIG. 5, the polarization controllers PC 1, PC 2output polarization compensated side carrier signals to circulator CIRC3, from which they are forwarded to splitter SS1. The splitter SS1 alsoshifts the phase of one of the output branches by 90 degrees relative toother branch. The output spectrum from SS1 is shown in FIG. 8 d. These 0degree and 90 degree phase shifted carriers are recombined in combinersC5 and C6, respectively, with the data bands output from splitter S7.In-phase (0 degree shifted) and quadrature (90 degree-shifted) signalspectrums out of outputs of respective combiners C5 and C6 are shown inFIG. 8 e and FIG. 8 f. As can be discerned, in each spectrum, a sidecarrier is positioned in the center of a data band. Each side of thespectra is equivalent to a spectrum generated by a conventional homodynesystem in which the local oscillator frequency is matched to the centerfrequency of the data band. Furthermore, as in conventional homodynereception, the power of the central carrier frequency is boostedrelative to the data portion in order to the improve signal-to-noiseratio of the detected signal. The side carrier that has been shifted 0degrees can be used to detect the in-phase (I) 10 Gb/s data channelsfrom the transmitter (channels 1, 3) and the side carrier that has beenshifted 90 degrees can be used to detect the quadrature (90 degreeshifted) 10 Gb/s data channels (channels 2, 4).

The combined signal from C5 is sent through optical amplifier EDFA6 andthe combined signal from C6 is sent through optical amplifier EDFA7 tofinal 50 GHz spaced demultiplexers DWDM 3 and DWDM 4. Each of thedemultiplexers DWDM 3, DWDM 4 separate the data bands and side carriersin adjacent channels for electro-optic conversion at photodiodes PD 1,PD 2 and PD 3, PD 4 respectively. In this manner 10 GB/s data channels 1and 3 are separated in DWDM 3 and 10 Gb/s channels 2 and 4 are separatedin DWDM 4, resulting in the output of four separate 10 Gb/s datasignals.

In an implementation of the receiver according to the present invention,low-bandwidth photodiodes can be placed at reflective ends ofpolarization controllers in each leg of WDM demultiplexer to providemonitor outputs proportional to fluctuations in each of carriers, forexample caused by cross phase modulation (XPM). Since the respective 10Gb/s data channels corresponding to the side carriers generallyfluctuate in sympathy, the effect of carrier fluctuation can be removedif the monitor output fluctuations are subtracted from the outputs ofthe respective received 10 Gb/s output channels.

After the converted data signals are further processed throughtrans-impedance amplifiers TIA1, TIA2, TIA 3, TIA 4 and low pass filtersLPF1, LPF2, LPF3, LPF4, they are input to a chromatic dispersioncompensation stage shown schematically in FIG. 9. It is noted in thiscontext that the dispersion compensation stage can equally beimplemented at the quadrature data modulators on the transmitter sideinstead of, or in addition to, implementation at the receiver. Theeffects of fiber-induced chromatic dispersion on quadrature-modulatedsinusoidal data signals can be described by the following matrixequation:

$\begin{matrix}{\begin{bmatrix}{{I\_ out}( {D,L,f} )} \\{{Q\_ out}( {D,L,f} )}\end{bmatrix} = {\begin{bmatrix}{\cos\;{{\phi 1}( {D,L,f} )}} & {\sin\;\phi\; 1( {D,L,f} )} \\{{- \sin}\;\phi\; 1( {D,L,f} )} & {\cos\;\phi\; 1( {D,L,f} )}\end{bmatrix}\begin{bmatrix}{{I\_ in}(f)} \\{{Q\_ in}(f)}\end{bmatrix}}} & (1)\end{matrix}$where I_out(f) and Q_out(f) are frequency domain representations ofoutput I amd Q signals, which are modified from frequency domainrepresentations of input I and Q signals, I_in(f) and Q_in(f), by thedispersion matrix, for which

$\begin{matrix}{{\Phi\; 1( {D,L,f} )} = {D \cdot L \cdot \frac{0.8}{4 \cdot \pi} \cdot 10^{- 26} \cdot ( {2\pi\; f} )^{2}}} & (2)\end{matrix}$D denotes the fiber dispersion in units of ps/nm*km, L stands for fiberlength in meters and f stands for frequency in Hz.

The dispersion matrix can be interpreted as a transfer function whichapplies a clockwise rotation angle that is proportional to the square ofthe frequency of the transmitted sinusoid. To counter the dispersioneffect, it is feasible to apply an inverse transfer function, which canbe interpreted as a counterclockwise rotation, also proportional to thesquare of the frequency. This counter-dispersion, or correction functionmay be described by the following matrix equation:

$\begin{matrix}{{{disp\_ corr}( {D,L,F} )} = \begin{bmatrix}{\cos\;( {{\phi 1}( {D,L,f} )} )} & {{- \sin}\;( {\phi\; 1( {D,L,f} )} )} \\{\sin( \;{\phi\; 1( {D,L,f} )} )} & {\cos( \;{\phi\; 1( {D,L,f} )} )}\end{bmatrix}} & (3)\end{matrix}$

Therefore to correct the I and Q data signal for the effects ofchromatic dispersion, the correction function is applied to the I and Qinput signals (again, either at the transmitter or at the receiver, asis shown). Multiplying the correction function by the input signalsyields:I_out=cos φ1(D, L, f)·I_in−sin φ1(D, L, f)·Q_inQ_out=sin φ1(D, L, f)·I_in+cos φ1(D, L, f)·Q_in  (4)

From equation (4), it is clear that dispersion compensation can beobtained by modifying the input I and Q data signals with an appropriatetransfer function and then combining the modified signal. An embodimentof a dispersion correction circuit that performs these operations isshown in FIG. 9. As shown, the I input signal is input to a splitterS10, from which an upper branch is delivered to amplifier A1 and a lowerbranch is delivered to an amplifier A2 in order to boost the signal. Theupper branch is transmitted to a COS1 circuit which applies the cosineportion of the dispersion correction function cos φ1 (D,L,f) to theinput data signal as will be described further below. The lower branchfrom the splitter S10 is fed to a SIN1 circuit which applies thecomplementary sine portion of the dispersion correction function.

The Q data signal is concurrently input to splitter S11 and broken upinto an upper branch which is fed through amplifiers A3, and a lowerbranch which is delivered to inverting amplifier IA1 which, in additionto boosting the signal, also shifts the phase of the signal by 180degrees. The upper and lower branches are thereafter input to respectiveCOS2 and SIN2 circuits which perform the same functions as the COS1 andSIN1 circuits, respectively. As shown, the modified signal from the SIN1circuit, which is the product I_in times sin φ1 (D,L,f), is combinedwith the output from COS2, the product, Q_in times cos φ1 (D,L,f), atcombiner CMB 1. Comparison with equation (4), shows that the output ofcombiner CMB1 matches the desired Q_out output for dispersioncompensation. Similarly, the combination at CMB2, containing theproducts I_in times cos φ1 (D,L,f) and Q_in times −sin φ1 (D,L,f),matches the desired I_out output for dispersion compensation.

Furthermore, the TX ID pilot signal, which, as noted above, is modulatedonto the reference frequency +/−10-100 kHz, is received at thepolarization controlllers PC 1, PC 2 and converted to the RF domain atphotodetectors PD3 and PD4. The TX pilots may be coded by frequencymodulation or by another code modulation technique. The TX ID identifiesthe particular transmitter sending the signal, allowing information,such as the length of optical fiber between the transmitter and thereceiver (which is the same as the parameter, L, used in the dispersioncorrection function), to be extracted from the coded signal. Thisinformation is transmitted to the chromatic dispersion compensationstage where it is received by a chromatic dispersion module 250. Thechromatic dispersion module, in turn, is coupled to the SIN and COScircuits and causes adjustments to be made to the respective transferfunctions applied to the I and Q inputs in accordance with theinformation extracted from the TX ID.

According to an embodiment of the present invention, the SIN and COScircuits of FIG. 9 are implemented as microstrip circuits which uselayers or regions of copper deposited on a circuitboard having variouswidths and lengths, to adjust electromagnetic effects that modifysignals sent through the copper layers or regions. FIG. 10 a and FIG. 10b illustrate implementations of the sin φ1 (D,L,f) and cos φ1 (D,L,f)transfer functions respectively. As is known in the art, variouscombinations of linear strips, (denoted as MLIN), t-junctions (denotedas MTEE), and capacitive elements (cap1, cap2), again having variousadjustable lengths and widths are used to fine-tune the electromagneticwave effects in the copper regions to simulate the desired transferfunctions.

In the foregoing description, the method and system of the inventionhave been described with reference to a number of examples that are notto be considered limiting. Rather, it is to be understood and expectedthat variations in the principles of the method and apparatus hereindisclosed may be made by one skilled in the art and it is intended thatsuch modifications, changes, and/or substitutions are to be includedwithin the scope of the present invention as set forth in the appendedclaims. For example, although only a 10 Gbp/s digital baseband isdiscussed, the inventive principles herein may be applied to higher orlower data rates as the case may be.

What is claimed is:
 1. An optical data signal transmitter comprising: aMach-Zender modulator, the Mach-Zender modulator receiving an inputoptical signal and modulating a pair of side carriers onto the inputoptical signal, outputting an optical carrier signal; and at least twophase modulators, the at least two phase modulators receiving theoptical carrier signal and each generating an optical data signal bymodulating a pair of data signals onto at least two data bands; whereinthe data bands are spread in frequency when modulated onto the opticalcarrier signal, the spreading causing an amplitude of the optical datasignal to be reduced to zero during transitions between data symbols. 2.The transmitter of claim 1, further comprising: a second Mach-Zendermodulator, the second Mach-Zender modulator imprinting the input opticalsignal with an identification code to generate a TX ID, theidentification code including information concerning the transmitter;and a combiner, the combiner attaching the TX ID to the optical datasignal.
 3. The transmitter of claim 2, further comprising: apolarization transformer; wherein the at least two phase modulatorsgenerate a first optical data signal including data bands imprinted witha first pair of data channels, the first optical data signal having afirst polarization state, and a second optical data signal includingdata bands imprinted with a second pair of data channels, the secondoptical data signal having a first polarization state, the polarizationtransformer altering a polarization state of the first optical datasignal from the first polarization state to a second polarization state.4. The transmitter of claim 3, further comprising a dense wave divisionmultiplexing unit; and means for separating the first pair of datachannels from the second pair of data channels based upon differingpolarization states of the first and second optical data signals;wherein the first and second pairs of data channels are input toseparate channels of the dense wave division multiplexing unit.
 5. Thetransmitter of claim 1, wherein the pair of side carriers is modulatedonto the input optical signal at both above and below a referencefrequency of the input optical signal.
 6. The transmitter of claim 5,wherein a first side carrier of the pair of side carriers is modulatedonto the input optical signal at 30 Ghz above and below the referencefrequency, and a second side carrier of the pair of side carriers ismodulated onto the input optical signal at 20 Ghz above and below thereference frequency.
 7. An apparatus for compensating a quadraturemodulated optical data signal for effects of chromatic dispersionoccurring during transmission over optical fiber, comprising: a receiverconfigured to separate in-phase and quadrature components of the opticaldata signal; a converter configured to optoelectrically convert thein-phase and quadrature components of the optical data signal intoin-phase and quadrature data signals; a modifier configured to apply acorrective function to the in-phase and quadrature data signals, thecorrective function modifying the in-phase and quadrature data signalsin a manner that counteracts effects of chromatic dispersion on thein-phase and quadrature components of the optical data signal.
 8. Theapparatus of claim 7, wherein the corrective function is a function of acoefficient of fiber dispersion, a length of the optical fiber, andfrequency of the optical data signal.
 9. An apparatus for correcting aquadrature modulated optical data signal for effects of chromaticdispersion, comprising: a receiver configured to derive in-phase andquadrature data signals via a homodyne reception system; and a modifierconfigured to apply a corrective function to the in-phase and quadraturedata signals, the corrective function modifying the in-phase andquadrature data signals in a manner that counteracts effects ofchromatic dispersion on the in-phase and quadrature components of theoptical data signal.
 10. An apparatus for reducing the transmitted powerof a quadrature modulated optical data signal, comprising: a transmitterconfigured to provide a quadrature modulated optical data signal, thetransmitter further configured to, during all transitional states of thequadrature modulated optical data signal in which data symbols canchange in value, reduce the power to zero such that transmitted powerdecreases to zero at approximately a mid point of each of thetransitional states, where data signals are spread out by approximatelyfifty percent in the frequency domain equivalent to a multiplication bya sine wave at half the data rate such that each symbol returns to zeroat approximately a mid-point of the transitional states.
 11. Anapparatus for reducing the transmitted power of a quadrature modulatedoptical data signal, comprising: a transmitter configured to provide aquadrature modulated optical data, the transmitter further configuredto, during all transitional states of the quadrature modulated opticaldata signal in which data symbols can change in value, reduce the powerto zero such that transmitted power decreases to zero at approximately amid point of each of the transitional states, the transmitter furtherconfigured to combine the quadrature modulated optical data signal witha side carrier, and the transmitter further configured to transmit theside carrier with the quadrature modulated optical data signal.
 12. Theapparatus of claim 11, wherein the transmitted power is independent of adata pattern of the quadrature modulate optical data signal.
 13. Amethod of reducing the transmitted power of a quadrature modulatedoptical data signal, comprises the steps of: providing a quadraturemodulated optical data signal by a transmitter; during all transitionalstates of the quadrature modulated optical data signal in which datasymbols can change in value, reducing, by the transmitter, the power tozero such that transmitted power decreases to zero at approximately amid point of each of the transitional states; and combining thequadrature modulated optical data signal with a side carrier andtransmitting the side carrier with the quadrature modulated optical datasignal.
 14. The method of claim 13, wherein the transmitted power isindependent of a data pattern of the quadrature modulate optical datasignal.